This application discloses an invention that is related, generally and in various embodiments, to a system and method for reducing switching losses in a high frequency multi-cell power supply.
In certain applications, multi-cell power supplies utilize modular power cells to process power between a source and a load. For example, FIG. 1 illustrates various embodiments of a power supply (e.g., an AC motor drive) having nine such power cells. The power cells in FIG. 1 are represented by a block having input terminals A, B, and C, and output terminals T1 and T2. In FIG. 1, a transformer or other multi-winding device 110 receives three-phase, medium-voltage power at its primary winding 112, and delivers power to a load 130 such as a three-phase AC motor via an array of single-phase inverters (also referred to as power cells). Each phase of the power supply output is fed by a group of series-connected power cells, called herein a “phase-group”.
The transformer 110 includes primary windings 112 that excite a number of secondary windings 114-122. Although primary windings 112 are illustrated as having a star configuration, a mesh configuration is also possible. Further, although secondary windings 114-122 are illustrated as having a delta or an extended-delta configuration, other configurations of windings may be used as described in U.S. Pat. No. 5,625,545 to Hammond, the disclosure of which is incorporated herein by reference in its entirety.
Any number of ranks of power cells are connected between the transformer 110 and the load 130. A “rank” in the context of FIG. 1 is considered to be a three-phase set, or a group of three power cells established across each of the three phases of the power delivery system. Referring to FIG. 1, rank 150 includes power cells 151-153, rank 160 includes power cells 161-163, and rank 170 includes power cells 171-173. A master control system 195 sends command signals to local controls in each cell over fiber optics or another wired or wireless communications medium 190. It should be noted that the number of cells per phase depicted in FIG. 1 is exemplary, and more than or less than three ranks may be possible in various embodiments.
In the example of FIG. 1 there is a separate secondary winding for each power cell. However, the number of power cells and/or secondary windings illustrated in FIG. 1 is merely exemplary, and other numbers are possible. The secondary windings in each rank may have the same phase angle, which may differ from the phase angle of all the other ranks. For applications in which all the cells carry an equal share of the load power, this arrangement causes many of the harmonics in the cell input currents to cancel in the transformer 110, so that they are not passed through to the primary currents.
FIG. 2 illustrates various embodiments of a power cell 210 which is representative of various embodiments of the power cells of FIG. 1. The power cell 210 includes a three-phase diode-bridge rectifier 212, one or more direct current (DC) capacitors 214, and an H-bridge inverter 216. The rectifier 212 converts the alternating current (AC) voltage received at cell input 218 (i.e., at input terminals A, B and C) to a substantially constant DC voltage that is supported by each capacitor 214 that is connected across the output of the rectifier 212. The output stage of the power cell 210 includes an 11-bridge inverter 216 which includes two poles, a left pole and a right pole, each with two switching devices 217, which in this example are insulated gate bipolar transistors (IGBTs). The inverter 216 transforms the DC voltage across the DC capacitors 214 to an AC voltage at the cell output 220 (i.e., across output terminals T1 and T2), often by using pulse-width modulation (PWM) of the semiconductor devices in the H-bridge inverter 216.
As shown in FIG. 2, the power cell 210 may also include fuses 230 connected between the cell input 218 and tile rectifier 212. The fuses 230 may operate to help protect the power cell 210 in the event of a short-circuit failure. According to other embodiments, the power cell 210 may be identical to or similar to those described in U.S. Pat. No. 5,986,909 or U.S. Pat. No. 6,222,284 to Hammond and Aiello, the disclosures of which are incorporated herein by reference in their entirety.
FIG. 3 illustrates exemplary waveforms associated with various embodiments of a power supply controlled by PWM. The power supply includes six power cells per phase, but is otherwise similar to the power supply of FIG. 1. The waveforms show a reference signal 302, a carrier signal 304, a voltage 306 which is the sum of voltages from six power cells in phase A, and a load voltage 308 from phase A to neutral.
Referring to FIGS. 2 and 3, the reference signal 302 represents the desired output voltage for one pole of an H-bridge inverter 216 in a power cell. The carrier signal 304 is a symmetrical triangular waveform oscillating at the desired switching frequency. The reference signal 302 may be compared with the carrier signal 304 to control the switching of one pole of the H-bridge inverter 216. When the reference signal 302 is greater than the carrier signal 304, the pole is switched to the positive DC voltage from capacitors 214, otherwise the pole is switched to the negative DC voltage from capacitors 214. For the other pole of the H-bridge inverter 216, the desired voltage is the inverse of the same reference signal. Therefore, the inverse of the reference signal may be compared with the same carrier signal (or vice-versa) to control the other pole. The other cells in the same phase group may use the same reference signal, and time-displaced replicas of the carrier signal. The sum 306 of the output voltages of all the cells in the phase-group is shown in FIG. 3. The other two phase-groups use the same set of carriers, with replicas of the reference waveform that are displaced in phase by ±120°. Therefore, the other two phase-groups produce similar sum voltages, which are also displaced in phase by ±120°. These three sum voltages give rise to three line-to-neutral voltages on the load, one of which 308 is shown in FIG. 3. This PWM method results in all of the cells carrying an equal share of the load power, thus allowing many harmonics in the cell input currents to cancel in the transformer. Additional details of this PWM method may be found, for example, in U.S. Pat. No. 5,625,545.
The example of FIG. 3 shows a carrier signal 304 that is oscillating at a desired switching frequency that is ten times the frequency of the reference signal. In many motor-drive applications the maximum desired output frequency is 60 hertz. Thus, with respect to FIG. 3, if the maximum desired output frequency is 60 hertz, the switching frequency in FIG. 3 is 600 hertz. Modern switching devices, such as IGBTs, can easily switch at 600 hertz without excessive switching losses.
FIG. 4 illustrates a plot of the frequency spectrum of the load voltage in FIG. 3. The vertical axis is scaled so that the fundamental (wanted) component 402 registers zero dB. FIG. 4 shows that the lowest harmonic (unwanted) component 404 which exceeds −40 dB (1% of the fundamental) is the 89th harmonic. If the fundamental (wanted) frequency is at 60 hertz, then the 89th harmonic will be at 5340 hertz. This large separation in frequency between the wanted and unwanted components is characteristic of PWM, when the switching frequency is significantly greater than the wanted frequency. Often, the load 130 includes significant series inductance (for example, an AC motor), and the high frequencies of the unwanted voltage components allows this inductance to suppress the resulting unwanted currents.
However, there are many applications where the wanted frequency is much greater than 60 hertz. For example, there is an emerging trend to connect a high-speed motor directly to a high-speed compressor or pump, without an intervening step-up gearbox. For such applications, the motor may be driven by a source of high-frequency power in order to spin at 5,000 RPM or more. For motors with more than two poles, the required frequency is even higher.
When the wanted frequency is increased to several hundred hertz, it becomes more difficult to extend the PWM method of FIG. 3 while still maintaining a switching frequency much higher than the wanted frequency. At a switching frequency of several thousand hertz, the switching losses may become the dominant losses in the power supply, the IGBTs may have to be operated below their nominal current rating, and the cost per kilowatt would increase
FIG. 5 illustrates exemplary waveforms associated with various embodiments of a power supply controlled by PWM. FIG. 5 is similar to FIG. 3, but is different in that the carrier signal 504 is oscillating at a switching frequency that is only four times the wanted frequency of the reference signal 502. In comparison to FIG. 3, it is clear that there are fewer steps per cycle in FIG. 5. FIG. 5 also shows the sum 506 of the output voltages of all of the cells in the phase group, along with a line-to-neutral voltage 508.
FIG. 6 illustrates a plot of the frequency spectrum of the load voltage in FIG. 5. The vertical axis is scaled so that the fundamental (wanted) component 602 registers zero dB. FIG. 6 shows that the lowest harmonic (unwanted) component 604 which exceeds −40 dB (1% of the fundamental) is the 17th harmonic. By reducing the ratio of switching-to-reference frequency by a factor of 2.5 (from ten in FIG. 3 to four in FIG. 5), the ratio of unwanted to wanted frequencies has been reduced by a factor of 5.24 (from 89 in FIG. 4 to 17 in FIG. 6). The amplitudes of the unwanted currents of FIG. 6 are increased by a similar factor over the amplitudes of the unwanted currents of FIG. 4.
Even with a switching frequency of only four times a wanted frequency of several hundred hertz, some derating of the IGBTs, and some increase in tile data transmission rate, may still be necessary in the prior art.